Figure 40.4 [11, 23, 29]. Figure 40.5(a) shows the signal spectrum (via the fast Fourier transform, or FFT), where the strong pilot signal is visible together with the 6 MHz signal sideband. Figure 40.5(b) shows the search results in terms of correlation versus search steps (per segment). When the code replica is the code for Field 1, the larger peak corresponds to Field 1 while the smaller peak corresponds to Field 2. Note that the separation between the two peaks is exactly 313 segments apart per signal spec.
Figure 40.6 shows the code and pilot tracking results. Figure 40.6(a) shows the correlation function over 7 lags, each spaced by ½ symbols. Due to the low update rate (only once every 313 segments) and jitter in symbol rate, the conventional three‐correlator structure (i.e. the early, prompt, and late correlators) may no longer be capable of maintaining lock, particularly during large motion‐induced transients.
Figure 40.6(b) shows the correlation versus frequency offset from the nominal value. Since the receiver is stationary, this large offset is largely due to transmitter and/or receiver clock drift errors. Figure 40.6(c) shows the correlation peaks over a field of 313 segments. It is this operation that detects the field sync segment and identifies the peak location.
Figure 40.5 Acquisition of DTV signals.
Figure 40.6(d) shows the in‐phase (real) and quadrature (imaginary) components of the prompt channel (complex). The quadrature component goes to near zero, while the in‐phase component holds most of the signal power. Due to an initial sign, the in‐phase is negative in this plot. The quadrature is not zero but is biased, likely due to the fact that the code used as the replica for correlation is not balanced. Figures 40.6(e) and (f) show the code error (in symbols) and symbol rate error (in symbols per seconds) from the DLL loop.
Figure 40.6(g) is the scatter plot of the quadrature versus in‐phase components of the incoming signal samples after low‐pass filtering around the nominal pilot frequency offset. After convergence of PLL, the in‐phase and quadrature components are shown in Figure 40.6(h), where the quadrature component is rendered close to zero while the in‐phase component maintains most of the signal power. Large variations of the in‐phase component are caused by the information content (data symbols of eight levels) carried by the DTV signal.
Figures 40.6(i) and (j) show the PLL frequency error and phase error, respectively, when the loop bandwidth is 5 Hz. Figures 40.6(k) and (l) show the PLL frequency error and phase error, respectively, when the loop bandwidth is 30 Hz. It is clear that with a wider bandwidth, the convergence is faster but the estimates are noisier. In contrast, a narrower bandwidth is less noisy but the transient is longer.
In recognition of the significant effects of mobile fading, ATSC has introduced the ATSC Mobile DTV Standard (A/153) for mobile and handheld users (ATSC‐M/H) [41]. It builds on the fixed reception ATSC‐8VSB (A/53) physical layer [37] to mitigate mobile fading so as to enable mobile DTV reception [42]. In addition to strong coding schemes, ATSC‐M/H incorporates longer and more frequent training sequences for effective channel equalization against severe multipath. Since the training sequences are transmitted in place of data segments, it sacrifices data throughput for mobile reception. Indeed, ATSC‐8VSB has only 0.3% of symbols for training whereas ATSC‐M/H now has 6%, a 20‐fold increase. Examples of experimental ATSC‐M/H signals can be found in [11].
The ATSC Standard A/53 [37] contains a provision for identification of DTV transmitters through the use of “RF watermarking.” The RF watermark signal is a spread‐spectrum signal, whose insertion level can be set, at any time for operation, from well below the normal noise floor of the host 8‐VSB transmitter (e.g. 30 dB below) up to higher levels only used in out‐of‐service testing. As a Kasami code sequence, the RF watermark signal is clocked at the symbol rate of the host 8‐VSB signal (10.76 MHz) and truncated to 65,104 symbols per cycle, which therefore repeats four times per data field. Serial data at a low rate (four symbols per host 8‐VSB data field) are modulated (phase inversion) on the RF watermark signal to permit separate data transmission for remote control and other purposes. The use of RF watermark signals (Kasami sequences) for timing and positioning is analyzed in [43].
Figure 40.6 Tracking of DTV‐8VSB field sync codes (a)–(f) and pilot signals (f)–(l).
40.2.2 Acquisition and Tracking of DVB‐T Signals for Timing and Ranging
The DVB‐T signals [44] share the same traits for use as SOOP for PNT as the ATSC‐8VSB signals discussed in Section 40.2.1. However, there are two major differences worth noting as far as PNT is concerned. First, the ATSC‐8VSB broadcasting can be viewed as a frequency division multiple access (FDMA) system with pulse amplitude modulation (PAM), wherein each DTV station transmits in its own frequency band which a receiver needs to tune to. In general, the ATSC‐8VSB stations are asynchronous, mostly operating on their own frequency and clock. From time to time, the DTV stations broadcast common network programming and may synchronize to the GPS time. In contrast, the DVB‐T can be used in a SFN in which all transmitters in the same SFN cell operate on the same frequency (efficient use of the spectrum) and are synchronized to the GPS time, thus being a synchronous network. A receiver therefore can receive signals originating from different transmitters on the same frequency band.
Second, the DVB‐T standard utilizes orthogonal frequency division multiplexing (OFDM) modulation as its air interface. The OFDM modulation has been adopted by many modern wireless communication systems such as Wi‐Fi 802.11 [45], 4G/LTE [46], and ultra‐wideband radar [47]. It offers high spectral efficiency due to the use of orthogonal subcarriers, which overlap but do not interfere, with properly chosen subcarrier spacing and pulse shaping. Since its bandwidth is small compared to the coherent bandwidth of the channel, each subcarrier is distorted by flat fading, which can be easily corrected using simple channel estimation techniques (e.g. one parameter). More importantly, a guard interval is inserted between successive OFDM symbols to avoid inter‐symbol interference (ISI); that is, there is no ISI if the maximum delayed version of a preceding symbol (multipath) does not cross over the guard interval into the subsequent symbol. In OFDM, the guard interval is used to transmit an exact copy of the end portion of an OFDM symbol waveform
